Onkyo SE200-PCI sound card windows 10 driver issues / disable immezio 3D effects

October 8, 2015, 12:35 am

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Apologies for the Google-friendly title, I'm hoping people experiencing the same problem will find their way here.

Onkyo download page: http://www.jp.onkyo.com/support/pcau...d/se200pci.htm

Current version is 5.60C, last update was 2012 to be compatible with Windows 8. Driver package 5.60C installs without issues on Windows 10 64 bit. The AudioDeck utility installs as well.

However, all is not well:

On installation, Immezio 3D effects are enabled. This locks the sample rate at 48 kHz. 44, 96, and 192 kHz cannot be selected. Deselecting the Immezio 3D effects prompts a reboot, but the Immezio 3D effects remain enabled after rebooting. The card is stuck at 48 kHz. 3D effects (which enables the DSP processing such as Qsounds, EAX, A3D) cannot be shut off.

I see three possible workarounds:

1. Find a command line switch, or edit the installation batch file to disable Immezio 3D on installation or startup.

2. Edit the registry to disable Immezio 3D effects.

3. Fix the part of the driver which is responsible for disabling Immezio 3D effects so that it does what it is supposed to. When radio button enable/disable is toggled, an executable called RegReboot.exe is called to handle the process. This presumably edits the registry settings and prompts for a reboot. It requires administrator privileges to run. I have attached it to this post along with the installation config file EnMixCPL. Maybe someone with more programming know-how than me can offer figure it out. I am fairly certain the problem is within RegReboot. The functions it is supposed to call to disable 3D effects are invalid under W10.

It's really frustrating as it's just this one little switch standing between me and being able to use my favorite soundcard in Windows 10.

P.S. Everyone I know with these cards (so far, four people) has had the same problem, or variants thereof. No one has managed to get the cards running properly in W10. Onkyo has told me they will not update the drivers for these cards to support Windows 10. So we are on our own.

update: VIA have released a Windows 10 driver for the VIA Vinyl VT1705, VT1708S, VT1802P, VT1802S, VT1828S, and VT2021 HD Audio codecs. (portal) The Onkyo card uses the VT1724 aka Envy24HT, however, which VIA have not supported past Windows 7 64 bit (with 5.60C, which Onkyo modified and made to work with 8)

Attached Files

RegReboot.zip (159.9 KB, 21 views)

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ASUS Xonar Essence STX soundcard

October 9, 2015, 3:46 pm

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https://www.asus.com/Essence-Hi-Fi-A...r_Essence_STX/

I admit I did not give ASUS the benefit of the doubt and seriously consider their Essence STX soundcard as a replacement for my Onkyo SE200-PCI. ASUS make nice motherboards, but unlike Onkyo have no previous expertise in high end audio.

I am happy to report - a bit late in the game, the card came out in 2009 - that they've done a really good job with it and the drivers for Windows 10, technically still in beta, work just fine.

*****

Asus updated the design recently to the STX II. The PCB has been redone, but the only visible change is the PCM1792A DAC has been moved towards the top of the card closer to the IV conversion op amps. An second LDO regulator IC, U34, empty on the STX, is now populated. A "TXCO" clock source is added next to the ASUS audio controller IC. The four film caps next to the output IC are replaced with WIMA brand. It's basically identical, so it is curious that ASUS bothered with a new PCB layout. It suggests to me that there was something in the original layout that needed to be fixed.

*****

Cool things about the ASUS STX:

TPA6120A2 headphone amplifier, with software programmable 0/+12/+18dB gain.

Relays (physical, I mean "click") to switch the output between line and headphone, rear and front panel.

DAC IV and filter stage op amps are socketed.

Power is connected through a 4 pin molex rather than the PCIe x1 bus.

PCM1792A DAC is definitely top class.

I haven't used it yet, but the ADC stage appears to be very high quality, so this card should be useful for recording my LPs.

Card comes with a full set of test measurements for line in and line out. Data for the headphone stage is curiously absent.

Driver control software utility looks like a Winamp skin from 1998 (sigh...) but it works fine.

*****

Sound quality. 1) Line output. Disappointing only in that it hints at greatness while falling short. It is musical and lively and pleasant, but its not the last word in either authority or refinement. Soundstage is frustratingly fuzzy and indistinct. 2) Headphone output. People have reported problems with low impedance headphones. My 300 ohm Sennheisers, at any rate, were driven well enough. Just a bit less in every respect than what is available from the line out into my Sapphire3.

That may sound like faint praise, but you have to take into account that I find many digital components straight-up unlistenable. To get where the STX is, which is a consistent 70-80% of where I would ideally want to be, with no outstanding flaws, is actually a pretty solid achievement.

And I do think you have to factor the convenience of an internal card with no additional boxes / clutter and the option to route the line and headphone output to the chassis front panel audio jack... as well as the relatively modest cost. All in all a solid value.

*****

There is a button "HF" (hi fi) that you are supposed to press on the control panel to bypass any DSP. With HF mode enabled, and 24/192, I have to say the STX has risen a notch or two in my estimation. There is pretty amazing ambient retrieval, but the soundstage is still indistinct. It is pretty, but lacks presence.

There is a very long thread on people's impressions of the STX over at head-fi, here. I figure I might as well try a few different op amps to see if it makes any difference.

I also plan to upgrade the four 2.7 nF film caps that are used for the LP filter around the buffer op amp to Wima FKP.

*****

2.7 nF caps replaced with WIMA FKP (FKP2D012701D00HA00). All op amps replaced with TI NE5532AP. This is my baseline reference, with trusty, work-well-in-anything 5532. Heck, even the PCM1792A datasheet specifically recommends the 5534 for the IV stage for optimal performance.

*****

Funny what familiarity brings. I know the NE5532/4 already quite well, and putting those op amps into the STX immediately made it feel more comfortable. The 5532 has a distinct "burr", a warm throatiness that I basically enjoy. In comparison the LM4562/JRC2114D set were way too lean.

*****

Comparing the headphone output and line output allows an indirect way to check the effect of the substitutions in the IV stage (both outputs) and LP filter stage (line out only). It suggests that most of the improvement actually came from substituting the JRC2114D for NE5532 for in the IV stage. The difference between the LM4562 and NE5532 in the filter stage is more subtle.

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B-board v2 line driver : development circuit

October 30, 2015, 5:49 am

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Original version is here.

I've been meaning to get around to updating this by folding in the improvements to the diamond buffer stage made during development of the Sapphire 3 headphone amplifier. Here is the first look of the bboard v2 under LTSpice.

I've gone back to simple emitter resistors on the input, running under much lower current to keep the input impedance high. The output is simplified to a basic Sziklai compound transistor pair with the bulk of the bias current running in the second transistor.

In terms of distortion, for line level output level, CCS loaded input has no advantage. I'll have to double-check PSRR and a few other things before signing off on this version though.

FYI only, not a production circuit.

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Sapphire3 bboard 10 mod.asc (4.5 KB, 7 views)

What are "Ground Boxes"?

January 23, 2016, 4:48 pm

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Ladies and gentlemen, please try to keep a straight face while browsing the following link.

http://www.entreq.com/products/ground-boxes-17667704

I'm not sure whether its possible to build a passive, permanent device that dissipates/neutralizes electrical charge. But the scientist part of me finds the claims that you can interesting.

Static electricity will eventually dissipate by attracting counter ions from the air. This happens more quickly if the humidity is high.

So seriously, if you just stuck a wire into a bucket of dirt, how much "earthing" would that actually provide? Is there any way to amplify that effect by using special materials or even passive electrical components?

Headphone amplifier ground. Where to make the chassis connection?

February 2, 2016, 12:11 am

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The discussion thread at the headphone forum is here, but I wanted to throw out the problem to the general blog-reading community here at diyaudio to see if anyone can nail this.

The earthed chassis (light blue) must connect to the circuit common i.e. "ground" (pale green). I do not know where the best place on the circuit ground is to tie that connection.

Suggestions please!

(COM and GND are completely equivalent pads on the circuit board, while IN- and OUT- also pads on the board but physically further away on the ground plane.)

****

Answer: as long as it connects at one point only, or the same point of both channels, it doesn't seem to matter at all. I have it connected at the ground tab of the headphone jack and that seems to be as good as anywhere.

****

The noise was in fact magnetic interference emanating from the transformers. Grounding layout changes / electrostatic shielding were ineffective, magnetic shielding was needed instead.

I bought a sheet of permalloy (mu metal) off ebay and wrapped a belt of the stuff around the transformers. This cut the interference drastically. RMAA-measured A-weighted S/N is now 97.7 dB.

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jFET phono stage | harmonic cancellation jujutsu

February 14, 2016, 11:45 pm

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There are lots of phono stage circuits floating around based on two jFET amplifier stages and a passive RIAA network. I'm not sure who did what first, but there's the Boozehound, LePacific, and of course Salas versions.

Setting aside concerns about the ripple rejection**, today I'd just like to focus on the distortion and noise of the circuit itself. The passive RIAA stage is a large obstacle. It attenuates the signal substantially at all but treble frequencies, and it generally presents a large series impedance - both of which tends to increase circuit noise.

The jFET themselves meanwhile are a fine balance between low current, low noise operation with high distortion, or running at high current, high noise, with low distortion. Circuit gain must be paid for meanwhile with added distortion since the two stage design struggles to manage 40 dB.

After spending some time in LTSpice with this, I realized it was the proverbial rock and hard place scenario. You can get something back though, because both stages are inverting. If the signal is the same magnitude and the loading is constant, even harmonic distortion will cancel going through both stages. As the signal attenuation/loading of the RIAA filter varies with frequency, you can only set this up to work perfectly at one frequency, but if 1 kHz is chosen it should be fairly close over most of the midrange.

The result of an afternoon's experiment is shown below, along with the LTspice circuit. It only manages about 34 dB gain, but I found bypassing the sources of the jFET amplifiers added too much distortion. Buffering both stages with followers allows more freedom to set the RIAA filter impedance and output loads lower. In short, yes, the second harmonic appears to be lower in the output that at other points in the circuit, at or below the noise floor. While I'm skeptical that the total distortion can be reduced in this way, it's basically the same trick as a push-pull amplifier: by making the distortion symmetrical over both halves of the waveform only odd-order harmonics are permitted.

This is just a concept, not a finished circuit. Haven't given jFET selection or the power supply design any serious thought yet.

** power supply rejection for this kind of circuit is truly appalling, it is even positive below 1 kHz. You heard that correctly: The circuit actually amplifies power supply noise, with gains up to 26 dB. Good grief!

*** LTSpice file replaced with slightly revised version of the circuit, now called "Crystal P".

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Crystal P.asc (3.3 KB, 18 views)

Passive RIAA equalization network response calculator

February 16, 2016, 1:21 am

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This Excel (2013) worksheet will help you fine tune the values of the resistors and capacitors used in the passive RIAA network found in any number of two stage tube, op amp, and FET phono stage circuits.

Excel handles complex numbers well enough now that this job isn't particularly difficult, though for simplicity the DC blocking cap (Cc) is left out of the calculation.

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Passive RIAA equalization network response calculator.zip (377.1 KB, 13 views)

jFET passive phono preamplifier circuit idea

February 16, 2016, 6:27 am

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There are various tricks, like parallel input devices and active current sources, that I have avoided here in the interests of simplicity. If you want to go down that road, you can get an idea where it leads, here. Instead, the circuit below is basically a JFET version of my old 6DJ8 amp, here. A single JFET was getting me nowhere in terms of output impedance - around 10kohms! - so I moved to a compound stage buffering each amplifier with a source follower.

Noise and distortion figures look okay. The gain is only 30 dB. A bit low. The main trick is the PSRR, which is awful. The two stage circuit actually amplifies the power supply noise onto the output. So considerable effort must be put into the power supply regulation and filtering. I note that this is pretty much par for the course with this circuit topology where resistors are used instead of current sources on the JFET drains.

The circuit below leaves out the usual RC filter inserted between the power supply and the JFETs. That will be dealt with later, as needed, with the rest of the supply circuit.

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LTSpice filter simulation masterclass: 0 to -100dB in five easy steps!

February 18, 2016, 4:14 pm

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I've never put everything into a single LTSpice worksheet like this before: I find it fascinating. You can really pull apart a circuit to see what makes it tick, before solder ever hits the iron.

Power supply ripple, frequency response, gain, and crosstalk can be established. You can look at turn on and turn off transients, inrush currents, and conductance angle, and check peak currents in the filter capacitors. It's all there if you care to peek in and poke around.

I'm such a huge fan of LTSpice... :D

The only problem, really, is it is too perfect: all devices are perfectly matched, every part value is exact, and the temperature is always 25 C. Ground loops, wiring inductance, and thermal runaway do not exist. So no, of course there are no guarantees - but as a tool to get you 90% of the way there with the minimum of fuss and bother it is truly indispensable.

Actually I find the more experience you have the more useful LTSpice gets, because you know much better what to look for in the simulation. For the example shown, torture-testing the shunt regulator with a pulsed current load made it clear which configurations would remain stable and which ones would fly off the rails...

*****

OK, so in the present case I have the whole circuit in front of me and I need to configure the power supply filtering such that the ripple at the output is minimal.

The phono stage itself has very poor ripple rejection. The PSRR is at bass frequencies (<100 Hz) is negative 20 dB. That means 1 mV of 60 Hz hum on the supply rail feeding the phono preamp circuit will end up as 10 mV on the line output.

The goal is to have <10 microvolts of ripple on the output, at any frequency. This means the power supply filtering must manage to provide an voltage rail with under 1 microvolts ripple/noise at bass frequencies. For an input ripple of about 1 V with fundamental 120 Hz that requires 120 dB ripple rejection at 120 Hz.

By way of comparison, an LM317 with the adjust terminal bypassed manages about 64 dB.

So how to achieve 120 dB? Here's some things that won't work:

1. A shoebox sized capacitor bank. The inrush current will blow your diodes or transformer, and even if they survive the charging currents will soon destroy the capacitors.

2. Passive filter CRCRCRC etc. As the capacitor impedance increases at lower frequencies, to be efficient at 120 hz this filter network would take 10 minutes to stabilize to a constant voltage.

3. Active single regulation stage with very high feedback. It can be done. Harder than it looks to keep stable though.

I finally opted for a shunt regulator with 75 dB RR, and CRC filter stages on the input and before each stage of the amplifier.

The input CRC gives an additional 20 dB, while the amplifier CRC stages another 40 dB. 20+40+75 = 135 dB, but the ripple from the first stage is about 5-6 V subtracting 15 dB from the total, giving the desired target of 120 dB.

The largest filter caps are right next to the amplifier, an inversion of the normal scheme of things. However, this circuit requires exceptionally high filtering and putting those large 1000 uF capacitors values higher up the chain towards the diodes would have resulted in unacceptably high peak charging currents.

Generally speaking ripple rejection is hardest to achieve at low frequency. So if the desired result is obtained at the ripple fundamental of 100 or 120 Hz, performance at higher harmonics will be as good or better still without having to tweak further.

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Voltage Regulators for Line Level Audio. Part 11 : The Crystal M Shunt

February 19, 2016, 5:49 pm

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A while back I did a series of blogs on voltage regulators. Back with a new entry today: The Crystal M, configured here for 40 V DC output and a 25 mA load.

The circuit is based on two p-channel MOSFETs, the top one is a constant current source, the bottom one a constant voltage source. As the load current changes, the voltage source adjusts its current to balance.

It's lifted directly on the Salas shunt design (as reworked by me for my own jFET phono stage), but the circuit can also be considered a distant, DC-coupled relative of the Zen amp.

I trick, I discovered, to getting it to work nicely - the attached screencap shows it well-behaved while handling a full-swing output current pulse - is the source resistor R10. This resistance dials-down the current gain of the MOSFET, damping out the overshoot.

The ripple rejection is about 70 dB over the audio bandwidth. The output impedance is about 0.05 ohms over the same frequency range.

One thing about shunt regulators: they run hottest when there is no load attached. So always calculate the power dissipation under this condition, and be careful not to leave the supply powered up but disconnected to the load: it uses just as much electricity this way as when the load is connected.

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Crystal M.asc (2.9 KB, 106 views)

Chromecast Audio Output Noise

February 26, 2016, 4:11 am

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Measured at 24/96 with my Asus Xonar STX soundcard (~ -147 dB noise floor)

The Chromecast Audio output noise powered with the included USB wall wart supply is -130 dB at 1 kHz, rising gradually at lower frequencies and showing some switching power supply noise peaks at 4763 Hz and higher multiples, never exceeding about -120 dB.

This is respectable performance given its price point.

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CrystalFET Phono Stage

March 2, 2016, 9:55 pm

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Development thread here.

What is it:

A moving-magnet jfet-based two stage phono preamp, with passive equalization, around 35 dB midband gain, and a dedicated, on-board MOSFET-based shunt voltage regulator.

What is good about it?

It is convenient to build, with a low item count BOM, easy-to-source-and-substitute parts, through-hole components, and straightforward, single-sided 8x10 cm PCB design. It runs off the same 2x12 VAC transformer and bridge rectifier supply shared with my VSPS, Phonoclone, and Sapphire projects.

The RIAA frequency response is very accurate with only standard-value parts needed.

Finally, the shunt power supply regulation has been extensively simulated for stability and fitness-for-purpose. The voltage regulation and filtering stages exceed 120 dB PSRR at 120 Hz.

LTSpice file attached. A very early version of the BOM is also attached (excel xlsx / .zip).

* Boards will be ordered shortly. If there is interest I will get boards and parts together for 8-9 kits to be sold at cost. (~<$100 US I expect, transformers/diodes/chassis/hardware not included.)

** Final parts values will be adjusted once I have jfets in hand to confirm the parameters. I'm looking for V_gs0 around 1~2 V. From the datasheets 2N5484 2N5457 J202 and J113 all seem viable. The stereo circuit requires two pairs of well-matched jfets and another four jfets for the followers that do not have to be closed matched.

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	Crystal 02.asc (10.6 KB, 12 views)
	pcb-crystal-00-bom.zip (127.3 KB, 22 views)

Matching JFETs

March 8, 2016, 5:29 am

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with only two resistors, a 9 V battery, and a voltmeter...

Current-voltage relationship for a jfet device is a quadratic expression defined by just two parameters, the saturation current, I_dss, and the pinch-off voltage, which I'll call V_gs0.

I = I_dss (1-V/V_gs0)^2

In principle, therefore, to characterize the device all we need is two data points (I1, V1) and (I2, V2). We don't need to measure I_dss or V_gs0 directly.

Certainly if our main purpose is matching devices then measurement precision is more important than accuracy, and it may likely be more convenient to make measurements away from the extremes of the transfer function defined by I_dss and V_gs0.

So, all you need to do is connect the jfet device-under-test (DUT) as shown, and measure the voltages across two different source resistances. That's it. The excel worksheet computes the I_dss and V_gs0 values for you (or you can do it by hand, the formulas are provided.)

The math is a bit messy, but if you can solve a quadratic expression it's easy enough.

The biggest limitation is that the resistors must be chosen in advance to be roughly appropriate for the parameters of the DUT. The excel worksheet attached will computer a range of workable values for you based on the expected device parameters.

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two measurement jfet match.zip (68.2 KB, 3 views)

Supermatched JFETs

March 28, 2016, 10:10 pm

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Truth be told, for a self-biased jfet audio circuit like the CrystalFET the main reason we need to used matched jfets is to ensure that the signal gain is the same in both channels. The operating point of the amplifier stage (the voltages and currents) can be allowed to vary a little so long as the transconductance (aka voltage gain) is the same.

Now, yes, ideally you would find two jfets with identical saturation current and pinch off voltages, ensuring not just the same gain but also the same operating point. In practice though you are usually binning parts that are close to each other based on some reference parameter that you hope closely correlates with the signal gain. This can be saturation current, or beta, or what I've been using: "size". None of those metrics, however, are as good as the calculating the actual transconductance of the particular device in the circuit it is to be used in. And since I can do this**, I figured: why not?

It allows for devices with slightly different transfer curves but the same circuit gain to be paired together. It is an efficient and accurate way of obtaining close gain match between channels.

The only caveat is the DC operating point has to be pre-screened so that the circuit voltages and currents are not wildly different.

The table below shows 20 J113 which have been pre-screened for "size" (V_gs0 x sqrt(I_dss)) 6-7. I could obtain four pairs of fets matched to g_m 3% and four pairs matched to 1%, with only four devices left unmatched.

_____

** transconductance, g_m = 2 sqrt(I_dss x I_ds) / V_gs0

[the derivative of the transfer function at I_ds]

If the saturation current is a little larger and the source drain current is a little smaller, the transconductance can balance out to the same value.

The only "trick" in the process is calculating I_ds from V_gs0, I_dss, and R_source.

I_ds = I_dss (1 - V_gs/V_gs0)^2 [jfet transfer function]

V_gs = I_ds x R_source [ohms law]

so

I_ds = I_dss [1 - ( I_ds x R_source)/V_gs0]^2

rearrange to the form ax^2 + bx + c = 0 with x=I_ds and solve the quadratic expression using x = (-b +\- sqrt(b^2 - 4ac)/2a.

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Understanding the J113 JFET

April 14, 2016, 6:08 am

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"To fight the bug, we must understand the bug."
--SKY MARSHALL TEHAT MERU

The J113 datasheet (Fairchild) tells you the following important information,

1. The (absolute) maximum gate-source / gate-drain voltage is 35 V.

2. The gate-source cut-off voltage (V_gs0) varies between -0.5 and -3 V.

3. The minimum zero-gate voltage drain current (I_dss) is 2 mA.

4. (from Fig. 11) the transconductance for I_ds 1-10 mA is about 10 mS largely independent of V_gs0.

5. (from Fig. 14) the voltage noise rises at low frequency and decreases with drain current, but is about 2-4 nV/sqrtHz over most of the audio bandwidth.

*****

I bought 200 J113 off eBay, but my measurements were set back after I realized my test rig was oscillating. Fixed that, and can now say a few things in addition to the datasheet.

The first is that the transfer curve (measured at RT, V_ds 10 V) does not perfectly follow the ideal square law. Although the full curve cannot be perfectly described by a single pair of V_gs0 and I_dss values, any small section of it can be closely approximated. In the first plot below you can see that one part of the curve fits to V_gs0 1.7 V I_dss 19 mA, while another is described by V_gs0 1.6 V I_dss 24 mA. This is important to keep in mind when simulating or calculating circuit gain or DC operating points. I'm going to quote parameters which fit to the curve at around 1-2 mA, since that is where I propose to use the devices...

Most of the J113 have V_gs0 around 1.3~1.8 V, with 1.5 V being typical. The beta (I_dss/V_gs0^2) is indeed fairly constant at about 9~10m, which implies that as the pinch off voltage increases, the saturation voltage increases to match. In fact I_dss values nearly all fall on a straight line plotted against V_gs0. If you know V_gs0, therefore, you can reliably infer I_dss.

The points plotted in the last two graphs below are not random samples of J113, but rather 4 devices pre-selected to show a wide range of V_gs0 varying from 1.3 to 1.7 V and another 12 which were all binned to have V_gs0 between 1.5 and 1.6 V. This gives you an idea of both the range and the scatter without going and plotting all 200 devices. The datasheet V_gs0 range is conservative. V_gs0 below 1 V or above 2 V are extremely rare, and most are between 1.3 and 1.8 V.

*****

What we learned:

1. V_gs0 cluster around the center of the range given in the datasheet.
2. The datasheet minimum I_dss isn't a very useful metric. On the other hand, the relatively constant beta value gives a very good handle on the device.

The J113 has,

V_gs0 1.5 V +/-20%
I_dss 22 mA +/-40%
beta constant around 10m

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Bboard buffer 2.0 (yet another version "2")

May 13, 2016, 1:22 am

≫ Next: Voltage Regulators for Line Level Audio. Part 12 : The Foursquare Shunt-Source

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A little bit of the sapphire headphone amp output stage, and a little bit of the LSK489 application note of all things (scroll down to Fig 10.).

Yes, this is probably the third "version 2" of this line buffer I've posted... I keep tossing it out and starting over. This variant looks pretty good: though the transistor count is a little high for such a basic function the performance is definitely there.

The main innovation re. the sapphire circuit is to replace the bias set resistors with diodes made out of the Vbe of transistors Q9 and Q10. This generates more voltage than is ideal, but can be handled by using largish values for the emitter resistors R13 and R14. Since this is a line stage buffer and not a headphone amplifier the output impedance of about 30 ohms and the limited output current swing are not critical flaws. It will drive 600 ohms at 0 dB with 0.001% THD. The whole circuit draws just 150 mW. The input impedance is a very high ~15 Mohms simulated (is that right?), while the PSRR is a usefully high 70 dB 0-20 kHz.

*****

rev. 2.1 ties the collectors of Q1,2 to the emitters of Q7,8 as seen in the LSK489 app note schematic. The distortion decreases slightly, with no other changes to the PSRR etc. or downsides that I can see.

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	bboard buffer 20.asc (4.6 KB, 9 views)
	bboard buffer 21.asc (4.6 KB, 2 views)

Voltage Regulators for Line Level Audio. Part 12 : The Foursquare Shunt-Source

June 17, 2016, 6:44 am

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I'm not totally sure this would work as advertised, but I can't see any obvious reason why it would not...

It's pretty much the same circuit as I used in the CrystalFET, which started out in a previous blog post in the Voltage Regulators for Line Level Audio series, but here I've replaced the MOSFETs with bipolars. It is shown configured to deliver 20 mA @ 12 V, split supply. Enough to power an op amp phono stage for example, or a preamp, or the voltage gain stage of a headphone amplifier.

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foursquare shunt regulator full.asc (4.6 KB, 2 views)

VSPSX phono stage design idea

June 23, 2016, 5:48 pm

≫ Next: INs and OUTs of the Asus Xonar Essence STX PCIe computer sound card

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The original VSPS is getting up in years, and could do with a bit of an update.

The VSPSX adds the foursquare shunt-source voltage regulator and bboard v2.1 output buffer.

The LTSpice schematic below abbreviates the transformer, rectifiers, and filter capacitors as V1 and V2. Op amp OP27 is a placeholder.

My main concern would be fitting all this stuff on a 80x100 board. That and, well, I'm not totally convinced yet that the complexity vs. reward ratio makes its worthwhile.

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VSPSX.asc (12.7 KB, 28 views)

INs and OUTs of the Asus Xonar Essence STX PCIe computer sound card

June 26, 2016, 1:12 am

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Asus Xonar Essence STX, Audacity 2.1.2, VLC, Windows 10 [and DigiOnSound6 Express for 24 bit recording]

Purpose

To confirm the calibration of the sound card input and output gain. Also, to determine the relationship between the signal voltage, the recorded signal amplitude displayed in Audacity, and the signal peak and noise baseline levels in the FFT spectra.

Summary

* Setting the volume slider of the device output to 100 gives 1 V rms output for an amplitude 0.5 sine wave.
* Setting the volume slider of the device recording line input to 100 gives records a 1 V rms tone as an amplitude 0.5 sine wave, which is displayed in the frequency spectrum (FFT) as peak of magnitude 0 dB in Audacity when both channels are averaged.
* volume setting 100 for unity gain loopback
* 0.5 amplitude sine wave = 0 dB FFT = 1 V rms
* noise baseline in averaged stereo FFT is 3 dB lower than single channel measurement
* The measured performance of the sound card is excellent.


dB scale for voltage

"dB" is 20 * log (V_measure / V_reference)

I will be consistent and always refer the absolute dB scale to V_reference = 1 V rms (1.4 V amplitude, 2.8 V peak-to-peak).

"deebees" are easy once you get used to them. Each time the voltage is doubled the signal increases 6 dB. Each time the voltage is halved, the signal decreases by 6 dB. So +18 dB is 2x2x2=6 times increase. -12 dB is 1/2 x 1/2 1/4 or 4 times smaller. A 10 times increase adds +20 dB.

Card output, with respect to signal amplitude

The thing to pin down is the relation between the signal amplitude as viewed in Audacity and the actual output voltage from the sound card. Audacity displays a waveform on a scale from -1 to 1. The "1" is not a voltage but represents the maximum value on the digital scale.

To settle this, we make a 1 kHz test tone in Audacity with an amplitude of 1 and 0.5 and check the output with a rms-capable voltmeter (Fluke 87).

At 100 volume, the 1 kHz, amplitude 1 test tone output is ~2 V rms. This is 6 dB. An amplitude 0.5 test tone is ~1 V rms. This is ~0 dB.

So if we want a 0 dB (1 V rms) test signal, we should make an amplitude 0.5 sinewave in Audacity and set the main device volume in Windows to 100. The output is 2 V rms, 6 dB, at full scale (i.e. amplitude 1). This is the standard for a line level output on consumer audio devices.

Incidentally the volume scale is logarithmic. The measured attenuation (dB) for a 0.5 amplitude reference is as follows,

vol L (dB) R (dB)
100 0.57 0.42
90 -1.02 -1.17
80 -2.79 -2.95
70 -4.81 -4.96
60 -7.15 -7.29
50 -9.97 -10.12
40 -13.32 -13.47
30 -17.70 -17.84
20 -23.74 -23.89
10 -34.47 -34.66

(the sound card has a 0.2 dB channel imbalance, fairly common in my experience)

The default level position of the Asus driver software is 50, or -10 dB. This is probably because the default output is expected to be the built-in headphone amplifier rather than line level output to a preamp. The measurements confirm that 100 and not 50 is the correct setting for true line level (home audio standard) 2 V rms full scale output.

Reference input

Playing a 0.5 amplitude sinewave test tone in Audacity at 100 volume gives an output of 0 dB, but to record in Audacity I'd want to play the file back in another application, VLC in my case. VLC has its own volume, with a maximum of 125. At 100 the output is -6 dB. At 125 is still isn't 0 dB. I had to open the Windows Sound Mixer and turn the volume slider for VLC to 100 to get 0 dB. This is why test and measurement on Windows is no fun. :mad:

At this point, I could record the amplitude 0.5 test tone sinewave .wav file as an amplitude 0.5 .wav file in Audacity. The input volume level was set at 100. I'm not going to mess with the input level control any further since 100 gives unity gain and decreasing the line input level would only attenuate the recorded signal.

One additional concern: Audacity only records in 16 bit on Windows, even if you set the file format to 24 or 32 bit and you have a 24 bit sound card. It's a licensing thing. So to show you what actual 24 bit looks like I also recorded the test tone playback using DigiOnSound6, then opened the file in Audacity. Either way the amplitude of the recorded data is 0.5.

Then, the FFT spectrum is calculated in Audacity. 65k samples, Hanning window. The peak at 1 kHz has an amplitude of 0 dB, confirming that 0 dB = 1 V rms = 0.5 amplitude and the FFT dB scale in Audacity is correctly calibrated so that a signal in both channels gives the dB for the signal in either channel.

To confirm, I disconnected one channel. The peak in the FFT spectrum decreases to -6 dB. Uncorrelated noise adds geometrically, however, so the same math that averages the L-R signals will reduce the uncorrelated noise by 3 dB:

two signals, adds numerically

(1 + 1)/2 = 2/2 = 1 = 0 dB

two signals, adds numerically

(1 + 0)/2 = 1/2 = 0.5 = -6 dB

two noise generators, adding geometrically

(sqrt(1^2 + 1^2)/2 = sqrt(0.5) = 0.707 = -3 dB

RMAA loopback test

As a cross check, and to get some measurements for the card's performance overall, I opened Rightmark and connected the sound card inputs and outputs together (loopback). The levels are 100 input, 100 output, and this gives the correct level indicator in Rightmark (Displays -0.8, -0.9 dB).

The noise baseline close the Audacity FFT recorded by DigiOnSound. RMAA records in 24 bit. The shape and absolute values are slightly different though. Since RMAA calculates each channel separately, uncorrelated noise would be expected to be reported 3 dB higher in RMAA than in Audacity (see above) but instead RMAA shows a lower noise baseline than Audacity. RMAA apparently uses some digital filtering which may be the cause for this, I don't know.

Rightmark gives the sound card full marks. The A-weighted SNR is -117 dB, in agreement with the published specification for the input stage (-118 dB). (The output stage is -124 dB supposedly, but of course I can't measure better than the card's input stage.)

Notes

The Xonar card has socketed, user switchable op amps for the I/V conversion and line output stages. I use TI NE5532AP in all three positions.

A general comment about noise floor measurements: use the best cables you have otherwise you'll just as likely to be measuring the noise picked up by the signal on its loop-back trip as that of the electronics itself.

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RMAA report Xonar STX loop back.zip (29.2 KB, 6 views)

RC filter coupling capacitor calculator

July 15, 2016, 5:54 pm

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When I need to pick the right capacitor for a coupling capacitor, rather than working out the time constant or 3dB cutoff I just remember the mnemonic "0.1-220" (meaning 0.1 uF and 220 kohms) and adjust the ratio up/down for the resistance I happen to be looking at: 0.22-100, 1-22, 0.47-47.

This amounts to a time constant (t=RC) of 20 ms, and 3 dB cutoff of 7 Hz. The bass attenuation at 20 Hz is half a dB.

If there are several stages the attenuation of all these filters add up, so it can be a good idea to make the capacitance about twice as large. There is rarely any advantage making it much larger still.

Excel worksheet attached. It spits out all the numbers so you don't have to guess.

* calculating the attenuation involves complex numbers. Zr=R, Zc=-i/(2 pi f RC), attenuation (high pass) = | Zr / (Zr+Zc) |. In excel you can use IMSUM, IMDIV, and IMABS to do the complex math.

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coupling capacitor time constants.zip (7.1 KB, 0 views)